Single stage cyclic analog to digital converter with variable resolution

ABSTRACT

A converter ( 200 ) adapted to convert an analog input signal into a digital output signal includes an analog input terminal ( 205 ) for receiving the analog input signal, a Redundant Signed Digit (RSD) stage ( 210 ) coupled to the analog input terminal, and a digital section ( 220 ). The RSD stage is configured to receive the analog input signal at the analog input terminal, produce a first number of bits at a digital output from the analog input signal during a first half of a first clock cycle, provide a residual feedback signal of the analog input signal at the analog input terminal during a second half of the first clock cycle, and produce a second number of bits at the digital output from the residual feedback signal during a first half of a second clock cycle, the second number of bits less than the first number of bits.

TECHNICAL FIELD

This disclosure relates generally to Analog-to-Digital (A/D) converters,and more particularly to Redundant Signed Digit (RSD) A/D converters.

BACKGROUND

Advances in integrated circuit technology have enabled the developmentof complex “system-on-a-chip” ICs for a variety of applications such aswireless communications and digital cameras. Such applications areembodied in portable electronic devices for which low power and smallcircuit area are important design factors. Low power and low voltagecircuits are needed to decrease battery power requirements, which canallow for designs that use fewer or smaller batteries, which in turndecreases device size, weight, and operating temperature.

Such devices, however, receive analog input signals that are typicallyconverted to digital signals. Various conventional cyclic (algorithmic)A/D converters that achieve low power operation and high resolution in asmall area are known. For example, U.S. Pat. No. 6,535,157, hereinincorporated by reference, discloses a cyclic RSD A/D converter having asingle RSD stage followed by a digital logic section that performssynchronization and correction functions.

Referring to FIG. 1, a block diagram of a cyclic RSD A/D converter 100,such as the one disclosed in U.S. Pat. No. 6,535,157, is shown. The A/Dconverter 100 includes an analog section having a single RSD stage 110followed by digital section 120 having an alignment and synchronizationblock 130 and a correction block 140. An analog input signal (e.g.,voltage) is input to the first RSD stage 110 by way of a first switch112. The RSD stage 110 provides a digital output signal to the digitalsection 120. The RSD stage 110 also generates a residual voltage signal(VR), which is fed back by way of the first switch 112. The first switch112 is closed for the first clock cycle, in which the analog inputsignal is received, and then opened for the remaining number of clockcycles that it takes to complete converting the analog signal to adigital signal. The feedback loop of the RSD stage 110 is directlyconnected from the RSD stage output to the first switch 112. The numberof clock cycles to complete the A/D conversion depends on the number ofbits in the digital output signal. The digital bits output from the RSDstage 110 are provided to the digital section 120, where they arealigned, synchronized, and combined to provide a standard format binaryoutput code.

While the single stage solution of U.S. Pat. No. 6,535,157 provides alow power, high resolution, and high speed A/D converter 100, there isstill a need for a single stage A/D converter that uses less power andhas a decreased silicon area. Embodiments of the invention address theseand other disadvantages of the related art.

SUMMARY

According to an example embodiment, a converter that is adapted toconvert an analog input signal into a digital output signal includes ananalog input terminal for receiving the analog input signal and aRedundant Signed Digit (RSD) stage coupled to the analog input terminal.According to the example embodiment, the RSD stage is configured toreceive the analog input signal at the analog input terminal, produce atleast two digital bits at a digital output from the analog input signalduring a first half of a first clock cycle, provide a residual feedbacksignal of the analog input signal at the analog input terminal during asecond half of the first clock cycle, and produce a single digital bitat the digital output from the residual feedback signal during a firsthalf of a second clock cycle. According to the example embodiment, theconverter further includes a digital section coupled to the digitaloutput, the digital section configured to combine the at least twodigital bits and the single digital bit into the digital output signal.

According to another example embodiment, a cyclic Redundant Signed Digit(RSD) Analog to Digital (A/D) converter includes an input terminal forreceiving an analog input signal, a first switch connected between theinput terminal and a first node, the first switch operable to apply theanalog input signal to the first node, and a second switch connectedbetween the first node and a second node, the second switch operable toapply a residual voltage feedback signal to the first node, the firstswitch operable to be closed when the second switch is open, the secondswitch operable to be closed when the first switch is open. According tothe example embodiment, the RSD A/D converter further includes anoperational amplifier having an output terminal connected to the secondnode, the operational amplifier operable to generate the residualvoltage feedback signal and apply it to the second node, comparators,each comparator having a first input coupled to the first node and anoutput, each of the comparators operable to compare a selected one ofthe analog input signal and the residual voltage feedback signal to apredetermined voltage signal, and a logic circuit coupled to the outputsof the comparators, the logic circuit operable to generate a firstdigital output signal during a first clock phase of an A/D conversionand operable to generate a second digital output signal during a secondclock phase of the A/D conversion, the first digital output signal basedupon the outputs from a first set of the comparators, the second digitaloutput signal based upon the outputs from a second set of thecomparators.

According to another example embodiment, a method for converting ananalog input signal into a plurality of digital bits of a digital outputsignal includes the steps of receiving the analog input signal,producing at least two digital bits of the plurality of digital bitsfrom the analog input signal during a first half of a first clock cycle,and producing a first residual voltage from the analog input signalduring a second half of the first clock cycle. The method furtherincludes producing a single bit of the plurality of digital bits fromthe first residual voltage during a first half of a second clock cycle.

BRIEF DESCRIPTION OF THE DRAWINGS

The following detailed description of some example embodiments will bebetter understood when read in conjunction with the appended drawings.It should be understood, however, that example embodiments are notlimited to the precise arrangements and instrumentalities shown. In thedrawings, like numerals are used to indicate like elements throughout.Furthermore, other desirable features and characteristics will becomeapparent from the subsequent detailed description and the appendedclaims, taken in conjunction with the accompanying drawings and theforegoing technical field and background.

For simplicity and clarity of illustration, the drawing figuresillustrate the general manner of construction, and descriptions anddetails of well-known features and techniques may be omitted to avoidunnecessarily obscuring aspects of the illustrated embodiments.Additionally, elements in the drawings figures are not necessarily drawnto scale. For example, the dimensions of some of the elements or regionsin some of the figures may be exaggerated relative to other elements orregions of the same or other figures to help improve understanding ofthe example embodiments. In the drawings:

FIG. 1 is a simplified schematic block diagram illustrating aconventional single-stage RSD A/D converter;

FIG. 2 is a schematic block diagram of a single-stage RSD A/D converterin accordance with example embodiments;

FIG. 3 is a schematic block diagram of an example embodiment of thesingle multi-bit/single-bit RSD stage of FIG. 2;

FIG. 4 is a schematic circuit diagram illustrating an example sub-ADCthat may be used to implement the multi-bit/single-bit RSD stage of FIG.3;

FIG. 5 is a schematic circuit diagram illustrating an example MDAC thatmay be used to implement the multi-bit/single-bit RSD stage of FIG. 3;

FIG. 6 is an example timing diagram illustrating control signals thatmay be used to perform an example 10-bit A/D conversion process with thesub-ADC of FIG. 4 and the MDAC of FIG. 5;

FIG. 7 is a simplified circuit diagram illustrating the configuration ofthe sub-ADC of FIG. 4 and the MDAC of FIG. 5 during the first clockphase of the example 10-bit A/D conversion process;

FIG. 8 is a simplified circuit diagram illustrating the configuration ofthe sub-ADC of FIG. 4 and the MDAC of FIG. 5 during the second clockphase of the example 10-bit A/D conversion process;

FIG. 9 is a simplified circuit diagram illustrating the configuration ofthe sub-ADC of FIG. 4 and the MDAC of FIG. 5 during the third clockphase of the example 10-bit A/D conversion process;

FIG. 10 is a simplified circuit diagram illustrating the configurationof the sub-ADC of FIG. 4 and the MDAC of FIG. 5 during the fourth clockphase of the example 10-bit A/D conversion process;

FIG. 11 is a simplified circuit diagram illustrating the configurationof the sub-ADC of FIG. 4 and the MDAC of FIG. 5 during the fifth clockphase of the example 10-bit A/D conversion process;

FIG. 12 is a simplified circuit diagram illustrating the configurationof the sub-ADC of FIG. 4 and the MDAC of FIG. 5 during the sixth clockphase of the example 10-bit A/D conversion process;

FIG. 13 is a simplified circuit diagram illustrating the configurationof the sub-ADC of FIG. 4 and the MDAC of FIG. 5 during the seventh clockphase of the example 10-bit A/D conversion process;

FIG. 14 is a simplified circuit diagram illustrating the configurationof the sub-ADC of FIG. 4 and the MDAC of FIG. 5 during the eighth clockphase of the example 10-bit A/D conversion process;

FIG. 15 is a simplified circuit diagram illustrating the configurationof the sub-ADC of FIG. 4 and the MDAC of FIG. 5 during the ninth clockphase of the example 10-bit A/D conversion process; and

FIG. 16 is a flowchart illustrating example processes included in amethod of A/D conversion according to an example embodiment.

DETAILED DESCRIPTION OF EXAMPLE EMBODIMENTS

The detailed description set forth below in connection with the appendeddrawings is intended as a description of some of the exampleembodiments, and is not intended to completely describe all possibleembodiments. That is, there is no intention to be bound by any expressedor implied theory presented in the preceding technical field,background, or the following detailed description of exampleembodiments. It is to be understood that the same or equivalentfunctions may be accomplished by different embodiments.

The terms “first,” “second,” “third,” “fourth” and the like in thedescription and the claims, if any, may be used for distinguishingbetween similar elements and not necessarily for describing a particularsequential or chronological order. It is to be understood that the termsso used are interchangeable under appropriate circumstances such thatthe embodiments described herein are, for example, capable of use insequences other than those illustrated or otherwise described herein.Furthermore, the terms “comprise,” “include,” “have” and any variationsthereof, are intended to cover non-exclusive inclusions, such that aprocess, method, article, or apparatus that comprises, includes, or hasa list of elements is not necessarily limited to those elements, but mayinclude other elements not expressly listed or inherent to such process,method, article, or apparatus.

FIG. 2 is a schematic block diagram of a single-stage RSD A/D converterin accordance with an example embodiment. The A/D converter 200 includesa single multi-bit/single-bit RSD stage 210 and a digital section 220.The digital section 220 has an alignment and synchronization block 230and a correction block 240. An analog input signal (e.g., voltage) 205is presented to an input of the single multi-bit/single-bit RSD stage210 by way of a first switch 212. The RSD stage 210 provides a digitaloutput signal to the digital section 220. The RSD stage 210 alsogenerates a residual voltage signal (VR), which is fed back to the inputof the RSD stage by way of the first switch 212. The first switch 212 isclosed for the first cycle, in which the analog input signal 205 isreceived, and then opened for the remaining number of cycles that ittakes to complete converting the analog signal to a digital signal.Preferably, the feedback loop of the RSD stage 210 is directly connectedfrom the RSD stage output to the first switch 212, without anyintervening circuitry such as a comparator. The number of cycles tocomplete an A/D conversion of the analog input signal to a digitaloutput signal depends on the number of bits in the digital outputsignal. The digital bits output from the RSD stage 210 are provided tothe digital section 220, where they are aligned, synchronized, andcombined to provide a standard format binary output code.

The architecture of FIG. 2 is capable of achieving significantreductions in total capacitance, area, and power relative to thearchitecture of FIG. 1. This is because, according to exampleembodiments, the single multi-bit/single-bit RSD stage 210 is initiallyconfigured to have a resolution of at least 2.5 bits during a firstclock phase of the A/D conversion, then reconfigured to have aresolution of 1.5 bits during subsequent clock phases of the A/Dconversion.

FIG. 3 is a schematic block diagram of an example embodiment 300 of thesingle multi-bit/single-bit RSD stage of FIG. 2. The RSD stage 300includes the input terminal 205 at which the analog input signal (VIN)is applied and a first switch 305 that is used to selectively apply theanalog input signal (VIN) to the node 307. The RSD stage 300 alsoincludes a feedback switch 315 that is used to selectively apply aresidual voltage feedback signal (VR) to the node 307.

The RSD stage 300 further includes a first, second, third, fourth,fifth, and sixth comparators 302, 304, 306, 308, 310, and 312,respectively. Because the RSD stage 300 has six comparators, it canachieve a maximum resolution of 2.5 bits. Although the six comparatorconfiguration illustrated in FIG. 3 is preferred, alternativeembodiments may have more than six comparators. In other words,alternative embodiments may achieve resolutions that are greater than2.5 bits. Each of the comparators 302, 304, 306, 308, 310, and 312 has apositive input terminal that is connected to the node 307. Depending onthe state of the first switch 305 and the feedback switch 315, thepositive input terminals of the comparators 302, 304, 306, 308, 310, and312 receive either the analog input signal or the residual voltagefeedback signal. That is, a selected one of the analog input signal andthe residual voltage feedback signal is input to the positive inputterminals of the first through sixth comparators 302, 304, 306, 308,310, and 312 through the use of the switches 305 and 315. Preferably,the residual voltage feedback signal VR is provided to the comparators302, 304, 306, 308, 310, and 312 via a direct feedback signal path asshown in FIG. 3 (i.e., no intervening circuitry, such as a sample andhold circuit).

Each of the comparators 302, 304, 306, 308, 310, 312 also has a negativeinput terminal that receives a first, second, third, fourth, fifth, andsixth predetermined voltage signals, respectively (e.g., VREF1, VREF2,VREF3, VREF4, VREF5, and VREF6). Each of the first, second, third,fourth, fifth, and sixth comparators 302, 304, 306, 308, 310, and 312compare the signals applied to their respective input terminals togenerate a comparator output signal.

According to an example embodiment, the RSD stage 300 is configurablesuch that, during an A/D conversion process for an analog input signalthat occurs over a number of sequential clock phases, the values of thepredetermined voltage signals (VREF1, VREF2, VREF3, VREF4, VREF5, andVREF6) may be selectively changed for each one of the clock phases. Forexample, during a first clock phase of the analog to digital conversion,each of the first, second, third, fourth, fifth, and sixth predeterminedvoltage signals (VREF1, VREF2, VREF3, VREF4, VREF5, and VREF6) may eachbe set to a unique value. During second and subsequent clock phases ofthe analog to digital conversion, some or all of the first, second,third, fourth, fifth, and sixth predetermined voltage signals (VREF1,VREF2, VREF3, VREF4, VREF5, and VREF6) may be changed to have adifferent value then in a previous clock phase.

According to the example embodiment, during clock phases of the A/Dconversion after the first clock phase, the RSD stage 300 uses outputsfrom less than all of the comparators 302, 304, 306, 308, 310, and 312.In other words, for clock phases after the first clock phase, theresolution that is achieved from the single-bit/multi-bit RSD stage 300is reduced relative to the resolution of the first clock phase. Theseaspects of the example embodiment are described in greater detail below.

The outputs of the first, second, third, fourth, fifth, and sixthcomparators 302, 304, 306, 308, 310, and 312 are connected to a logiccircuit 320. During clock phases of an A/D conversion process, the logiccircuit 320 is capable of generating a digital output signal that isrepresentative of the selected one of either the analog input signal orthe residual voltage feedback signal. In an example embodiment, thelogic circuit 320 generates three raw digital bits (D0, D1, D2) as thedigital output signal during a clock phase of an A/D conversion processbased upon the output from all the comparators 302, 304, 306, 308, 310,312. In the example embodiment, the logic circuit 320 generates two rawdigital bits (D0, D1) as the digital output signal during another clockphase of the A/D conversion process based upon outputs from less thanall of the comparators 302, 304, 306, 308, 310, 312. In a preferredembodiment, the three digital bits (D0, D1, D2) are generated during thefirst clock phase of the A/D conversion process. The digital bitsgenerated during any clock phase of the A/D conversion are aligned andsynchronized in the digital section 220, and then combined with thedigital bit or bits from other clock phases of the A/D conversion toform a formatted binary output code.

During clock phases of the A/D conversion, the logic circuit 320 is alsocapable of generating a high switch control signal 333, a mid switchcontrol signal 353, and a low switch control signal 343 based upon atleast two of the output signals from the comparators 302, 304, 306, 308,310, and 312.

The single-bit/multi-bit RSD stage 300 additionally includes aprogrammable gain/summing element 325. The programmable gain/summingelement 325 receives as inputs the high switch control signal 333, themid switch control signal 353, the low switch control signal 343, theselected one of the analog input signal and the residual voltagefeedback signal from node 307, a first reference voltage VREFP, a secondreference voltage VREFM, and a ground voltage. The programmablegain/summing element 325 generates the residual voltage feedback signalVR. Although the actual transfer function associated with theprogrammable gain/summing element 325 will be dependent upon theparticular design, generally speaking, the residual voltage feedbacksignal VR may be thought of as a sum of two products. Depending on theparticular clock phase of the A/D conversion, the first product iseither the analog input signal or a previously generated value of theresidual voltage feedback signal, multiplied by a first gain factor. Thesecond product is a selected one of the reference voltages (VREFP,VREFM, or zero), multiplied by a second gain factor.

The feedback switch 315 is provided for selecting the residual voltagefeedback signal as an input to the programmable gain/summing element 325and the comparators 302, 304, 306, 308, 310, and 312. The feedbackswitch 315 is disposed between the output of the programmablegain/summing element 325 and the node 307. When the feedback switch 315is closed, the first switch 305 is open so that the residual voltagefeedback signal is input to the programmable gain/summing element 325and the comparators 302, 304, 306, 308, 310, and 312. When the firstswitch 305 is closed, the feedback switch 315 is open so that the analoginput signal is input to the programmable gain/summing element 325 andthe comparators 302, 304, 306, 308, 310, and 312. As discussed above,the first switch 305 is closed in a first clock cycle during theconversion of the analog input signal and the first switch 305 is openfor subsequent cycles of converting the analog input signal.

FIGS. 4 and 5 are schematic circuit diagrams that illustrate in furtherdetail the multi-bit/single-bit RSD stage 300 of FIG. 3 in accordancewith an example embodiment. FIG. 4 illustrates a sub-ADC 400 inaccordance with an example embodiment, while FIG. 5 illustrates aMultiplying Digital-to-Analog Converter (MDAC) 500 in accordance with anexample embodiment. The multi-bit/single-bit RSD stage 300 illustratedin FIG. 3 may be implemented with the sub-ADC 400 of FIG. 4 and the MDAC500 of FIG. 5.

Referring to FIG. 4, the sub-ADC 400 includes the input terminal 205that receives an analog input signal (VIN). The first switch 305 isdisposed between the input terminal 205 and a first node 405. The firstswitch 305 is operable to selectively apply the analog input signal tothe first node 405 when the first switch is closed. The feedback switch315 is disposed between the first node 405 and a second node 410. Thefeedback switch 315 is operable to selectively apply the residualvoltage feedback signal (VR) to the first node 405 when the feedbackswitch is closed. As was discussed above, when the first switch 305 isclosed, the feedback switch 315 is open and when the first switch 305 isopen, the feedback switch 315 is closed. The first switch 305 is closedduring a first clock phase of an A/D conversion process and the feedbackswitch 315 is closed during subsequent clock phases of the A/Dconversion process.

The sub-ADC 400 further includes first, second, third, fourth, fifth,and sixth comparators 302, 304, 306, 308, 310, and 312, respectively.The operation of the comparators 302, 304, 306, 308, 310, 312 is thesame as what was described above for FIG. 3. The sub-ADC 400 furtherincludes the logic circuit 320. The logic circuit 320 is connected tothe first through sixth comparators 302, 304, 306, 308, 310, 312 andreceives the output signals from the first through sixth comparators.

According to an example embodiment, during the first clock phase of anA/D conversion process, the logic circuit 320 generates three rawdigital bits (D0, D1, D2) based upon the output signals from each of thefirst through sixth comparators 302, 304, 306, 308, 310, 312. Accordingto the example embodiment, the logic circuit 320 generates two rawdigital bits (D0, D1) during one or more subsequent clock phases of theA/D conversion process, based upon the output signals from less than allof the comparators 302, 304, 306, 308, 310, 312. The logic circuit alsogenerates control signals (h, l, m), which are used to control someswitches of the MDAC 500. This will be explained in further detailbelow. The control signals h, l, m correspond to the high, low, and midswitch control signals 333, 343, and 353 of FIG. 3, respectively.

The programmable gain/summing element 325 of FIG. 3 may be implementedusing the MDAC 500 of FIG. 5. Referring to FIG. 5, the MDAC 500 includesan operational amplifier (op-amp) 555, capacitors 521, 523, 525, 527,and 581, and switches 502, 504, 506, 508, 512, 514, 522, 524, 532, 534,536, 542, 544, 546, 552, 554, 556, 562, 564, 572, 574, 582, 584, 586,588, 590, 592, and 594, all of which are arranged in the manner shown inFIG. 5. For completeness, it should be noted at this point that otherMDAC designs that implement the programmable gain/summing element 325 ofFIG. 3 may be conceived by those of skill in the art that arestructurally dissimilar to MDAC 500 but nevertheless accomplish the samefunction. For example, some or all of the switches shown in FIG. 5 couldbe implemented using transistors.

The circuit connections illustrated in FIG. 5 are as follows. The analoginput signal at node 501 is selectively connected to the nodes 511, 513,515, 517 by way of the switches 502, 504, 506, and 508, respectively.The residual voltage feedback signal at node 503 is selectivelyconnected to the nodes 511, 513, 515, 517 by the switches 512, 514, 522,and 524, respectively. The nodes 513, 515, and 517 are selectivelyconnected to the predetermined high reference voltage source (VREFP) bythe switches 532, 534, and 536, respectively. The nodes 513, 515, and517 are selectively connected to the predetermined low reference voltagesource (VREFM) by the switches 542, 544, and 546, respectively. Thenodes 513, 515, and 517 are selectively connected to the predeterminedzero voltage source by the switches 552, 554, and 556, respectively. Thenode 511 is selectively connected to the node 503 by the switch 562.Capacitor 521 is coupled between the nodes 517 and 533. Capacitor 523 iscoupled between the nodes 515 and 533. Capacitor 525 is coupled betweenthe nodes 513 and 531. Capacitor 527 is coupled between the nodes 511and 531. Node 531 is selectively connected to the predetermined zerovoltage by switch 590. Node 533 is selectively connected to thepredetermined zero voltage by switch 592. Node 531 is selectivelyconnected to node 533 by switch 582. Node 531 is selectively connectedto the negative input terminal of the op-amp 555 by switch 564, whilenode 533 is selectively connected to the negative input terminal of theop-amp by the switch 574. Node 515 is selectively connected to the node503 through switch 572. The negative input terminal of the op-amp 555 isselectively connected to the node 583 by the switch 586. The negativeinput terminal of the op-amp 555 is selectively connected to the node503 by the switch 594. The positive input terminal of the op-amp 555 istied to the predetermined zero voltage. The capacitor 581 is selectivelycoupled between the node 503 and the node 583 by the switch 588. Node583 is selectively coupled to the predetermined zero voltage by theswitch 584.

According to an example embodiment, during a clock phase of an A/Dconversion process the MDAC 500 is operable to produce a first gainfactor of four (4) for the analog input signal and a selected secondgain factor of zero, one, two, or three (0, 1, 2, or 3) for a selectedone of the reference voltages (VREFP, VREFM, or zero). According to theexample embodiment, during subsequent clock phases of the A/D conversionthe MDAC 500 is operable to produce a first gain factor of two (2) forthe residual voltage feedback signal and a selected second gain factorof zero or one (0 or 1) for a selected one of the reference voltages(VREFP, VREFM, or zero). It should be recalled that that the MDAC 500illustrated in FIG. 5 is but one possible implementation for theprogrammable gain/summing element 325 of FIG. 3. Alternative embodimentsmay implement the programmable gain/summing element 325 with differentMDAC designs, but the embodiments will nevertheless still be capable ofapplying different gain factors to the analog input signal, the residualvoltage feedback signal, and the selected reference voltage duringdifferent clock phases of an A/D conversion.

Switches 305 and 315 of FIG. 4, as well as the switches included in theMDAC 500 of FIG. 5, control the operation of the multi-bit/single-bitRSD stage 300. As will become apparent in the following paragraphs, someof the switches in the MDAC 500 are controlled using control signalsthat are derived from a common clock signal, while other switches arecontrolled by the high, low, and mid-switch control signals (h, l, m)that are generated by the logic circuit 320 of the sub-ADC 400. Thedetails associated with deriving one or several control signals from aclock signal is not explained in further detail here, as they are notcritical for an understanding of the example embodiments.

The switches that are included in the MDAC 500 of FIG. 5 are all listedin the left column of Table I, which appears below this paragraph.Control signals are listed in the right column of Table I. For eachswitch or group of switches appearing in the rows of the left column,the corresponding row in the right column contains the control signal orcontrol signals that determine the state of the switch or group ofswitches. The (OR) notation for switches 590 and 592 refers to thelogical OR function.

TABLE I Switch(es) Control Signal 502, 504, 506, 508 SWVIN 512, 514SWFB1 522, 524 SWFB2 532, 534, 536 h (from logic circuit 320) 542, 544,546 l (from logic circuit 320) 552, 554, 556 m (from logic circuit 320)562, 564 MFB1 572, 574 MFB2 582, 584 H_(O) 586 H_(E) 588 H_(SH) 590SWVIN (OR) SWFB1 592 SWVIN (OR) SWFB2 594 reset

FIG. 6 is an example timing diagram illustrating control signals thatmay be used to perform an example 10-bit A/D conversion using thesub-ADC 400 of FIG. 4 and the MDAC 500 of FIG. 5. FIG. 6 illustrates aclock signal, as well as control signals that were named in Table I thatare derived from the clock signal. In FIG. 6, one clock cycle is definedas the time between adjacent rising edges of the clock signal. A clockphase is each “up” or “down” period of the clock signal. Thus, FIG. 6illustrates control signals from Table II for ten sequential clockphases or five clock cycles.

Table II, which appears below this paragraph, illustrates the state ofall the switches that are controlled by the control signals of FIG. 6during each of the ten clock phases illustrated in FIG. 6. In Table II,an “X” indicates that the associated switch or switches are closed,while the absence of an entry indicates that the associated switch orswitches are open. Table II was derived using Table I and FIG. 6. Forexample, according to Table 1, the state of switch 592 is controlledbased upon the logical expression SWVIN OR SWFB2. FIG. 6 illustratesthat the control signal SWVIN or the control signal SWFB2 is at a logic“high” state during the first, fourth, sixth, and eighth clock phases.Consequently, Table II illustrates that switch 592 is closed during thefirst, fourth, sixth, and eighth clock phases. The state of the otherswitches may be derived in the same manner. Table II therefore providesa convenient way to summarize the state of switches that are illustratedin FIG. 5 during the ten clock phases illustrated in FIG. 6.

TABLE II Clock Phase Switch(es) 1 2 3 4 5 6 7 8 9 10 502, 504, X 506,508 512, 514 X X X 522, 524 X X X 562, 564 X X X 572, 574 X X X 582, 584X 586 X 588 X X 590 X X X X 592 X X X X

FIGS. 7-15 are simplified circuit diagrams illustrating theconfiguration of the sub-ADC 400 of FIG. 4 and the MDAC 500 of FIG. 5for the first nine clock phases of the example 10-bit A/D conversionusing the control signals illustrated in FIG. 6. The simplified circuitdiagrams of FIGS. 7-15 may be obtained using the status of the switchesduring each phase of the 10-bit A/D conversion as summarized in TableII. Thus, FIG. 7 corresponds to the first clock phase, FIG. 8corresponds to the second clock phase, FIG. 9 corresponds to the thirdclock phase, etc., up to FIG. 15, which corresponds to the ninth clockphase. A diagram corresponding to the tenth clock phase omitted because,as can be seen in Table II, all switches are open resulting in anuninteresting configuration for MDAC 500. In order to increase clarity,none of the switches illustrated in FIG. 5 are shown in the simplifiedcircuit diagrams of FIGS. 7-15, and any capacitor 521, 523, 525, 527,581 that is grounded on both sides during a particular clock phase isnot illustrated in the FIGURE corresponding to that clock phase. In theparagraphs that follow, the example 10-bit A/D conversion process asperformed by the sub-ADC 400 of FIG. 4 and the MDAC 500 of FIG. 5 willbe explained in greater detail.

FIG. 7 is a simplified circuit diagram illustrating the configuration ofthe sub-ADC 400 and the MDAC 500 during the first clock phase of theexample 10-bit A/D conversion process. During the first clock phase, theanalog input is sampled by the MDAC 500 and the sub-ADC 400, which usesoutputs from all of the comparators 302, 304, 306, 308, 310, and 312 togenerate three bits of raw digital data which are sent to the digitalsection 220 of FIG. 2. During the first clock phase, the reset signal isasserted to close the switch 594 of FIG. 5, which results in theresetting of op-amp 555. According to other embodiments, the op-amp 555may be reset during the first clock phase for any n-bit A/D conversionprocess.

FIG. 8 is a simplified circuit diagram illustrating the configuration ofthe sub-ADC 400 and the MDAC 500 during the second clock phase of theexample 10-bit A/D conversion process. During the second clock phase,the op-amp 555 generates the residual voltage feedback signal VR₁ basedupon the full sub-ADC 400 results from the previous first clock phase.As was explained above, VR₁ is generated using four as the first gainfactor for the analog input signal and using two as the second gainfactor for a selected one of the predetermined reference voltage sources(VREFP, VREFM, zero). The capacitors 521, 523, and 525 are tied toeither the predetermined high reference voltage source (VREFP), thepredetermined low reference voltage source (VREFM), or the predeterminedzero reference voltage based upon the high, low, and the mid switchcontrol signals (h, l, m). As was explained above, the high, low, andmid switch control signals h, l, m determine the state of the switches532, 534, 536, 542, 544, 546, 552, 554, 556 (FIG. 5) of the MDAC 500.During the second clock phase, the residual voltage feedback signal VR₁is sampled on to the capacitor 581. Note that during the second clockphase, the sub-ADC 400 does not generate a digital output signal of oneor more raw digital bits. According to other embodiments, the sub-ADC400 does not generate a digital output signal during the second clockphase for any n-bit A/D conversion process.

FIG. 9 is a simplified circuit diagram illustrating the configuration ofthe sub-ADC 400 and the MDAC 500 during the third clock phase of theexample 10-bit A/D conversion process. During the third clock phase, theresidual voltage feedback signal VR₁ generated during the previoussecond clock phase is held and sampled on to the capacitors 525 and 527as well as sampled by the comparators 302 and 304 of the sub-ADC 400.During the third clock phase, the first predetermined voltage signal(VREF1) may be a predetermined high voltage (VH) that is applied to thenegative input terminal of the first comparator 302. During the thirdclock phase, the second predetermined voltage signal (VREF2) may be apredetermined low voltage (VL) that is applied to the negative inputterminal of the second comparator 304. The actual voltage values for VHand VL are a function of process technology since that may limit powersupply voltages. However, in one example embodiment, VH is about 1.5Volts (V) and more preferably about 1.475 V, while VL is about 1.2 V andmore preferably about 1.225 V. Based upon the output signals from thecomparators 302, 304, the logic circuit 320 of the sub-ADC 400 maygenerate new values for the high, low, and mid switch control signals(h, l, m). The sub-ADC 400 also generates two raw digital bits at theend of the third clock phase, which are sent to the digital section 220of FIG. 2. It will be appreciated by those of ordinary skill that theresolution achieved by the single-bit/multi-bit stage 300 during thisclock phase is only 1.5 bits, since only two comparators 302, 304 of thesub-ADC 400 are used.

FIG. 10 is a simplified circuit diagram illustrating the configurationof the sub-ADC 400 and the MDAC 500 during the fourth clock phase of theexample 10-bit A/D conversion process. During the fourth clock phase,the op-amp 555 generates a new residual voltage feedback signal VR₂ fromthe previous residual voltage feedback signal VR₁ based on the controlsignals (h, l, m) from the previous third clock phase. The residualvoltage feedback signal VR₂ is held and sampled on to the capacitors 521and 523 as well as sampled by the comparators 306, 308 of the sub-ADC400. During the fourth clock phase, the third predetermined voltagesignal (VREF3) may be the predetermined high voltage (VH) that isapplied to the negative input terminal of the third comparator 306.During the fourth clock phase, the fourth predetermined voltage signal(VREF4) may be the predetermined low voltage (VL) that is applied to thenegative input terminal of the fourth comparator 308. Based upon theoutput signals from the comparators 306, 308, the sub-ADC 400 maygenerate new values for the high, low, and mid-switch control signals(h, l, and m). The sub-ADC 400 also generates two raw digital bits atthe end of the fourth clock phase, which are sent to the digital section220 of FIG. 2.

FIG. 11 is a simplified circuit diagram illustrating the configurationof the sub-ADC 400 and the MDAC 500 during the fifth clock phase of theexample 10-bit A/D conversion process. During the fifth clock phase, theop-amp 555 generates a new residual voltage feedback signal VR₃ from theprevious residual voltage feedback signal VR₂ based on the controlsignals (h, l, m) from the previous fourth clock phase. The residualvoltage feedback signal VR₃ is held and sampled on to the capacitors 525and 527 as well as sampled by the comparators 310, 312 of the sub-ADC400. During the fifth clock phase, the fifth predetermined voltagesignal (VREF5) may be the predetermined high voltage (VH) that isapplied to the negative input terminal of the fifth comparator 310.During the fifth clock phase, the sixth predetermined voltage signal(VREF6) may be the predetermined low voltage (VL) that is applied to thenegative input terminal of the sixth comparator 312. Based upon theoutput signals from the comparators 310, 312, the sub-ADC 400 maygenerate new values for the high, low, and mid-switch control signals h,l, and m. The sub-ADC 400 also generates two raw digital bits at the endof the fifth clock phase, which are sent to the digital section 220 ofFIG. 2.

FIG. 12 is a simplified circuit diagram illustrating the configurationof the sub-ADC 400 and the MDAC 500 during the sixth clock phase of theexample 10-bit A/D conversion process. During the sixth clock phase, theop-amp 555 generates a new residual voltage feedback signal VR₄ from theprevious residual voltage feedback signal VR₃ based on the controlsignals (h, l, m) from the previous fifth clock phase. The residualvoltage feedback signal VR₄ is held and sampled on to the capacitors 521and 523 as well as sampled by the comparators 302, 304 of the sub-ADC400. During the sixth clock phase, the first predetermined voltagesignal (VREF1) may be the predetermined high voltage (VH) that isapplied to the negative input terminal of the first comparator 302.During the sixth clock phase, the second predetermined voltage signal(VREF2) may be the predetermined low voltage (VL) that is applied to thenegative input terminal of the sixth comparator 304. Based upon theoutput signals from the comparators 302, 304 the sub-ADC 400 maygenerate new values for the high, low, and mid-switch control signals(h, l, and m). The sub-ADC 400 also generates two raw digital bits atthe end of the sixth clock phase, which are sent to the digital section220 of FIG. 2.

FIG. 13 is a simplified circuit diagram illustrating the configurationof the sub-ADC 400 and the MDAC 500 during the seventh clock phase ofthe example 10-bit A/D conversion process. During the seventh clockphase, the op-amp 555 generates a new residual voltage feedback signalVR₅ from the previous residual voltage feedback signal VR₄ based on thecontrol signals (h, l, m) from the previous sixth clock phase. Theresidual voltage feedback signal VR₅ is held and sampled on to thecapacitors 525 and 527 as well as sampled by the comparators 306, 308 ofthe sub-ADC 400. During the seventh clock phase, the third predeterminedvoltage signal (VREF3) may be the predetermined high voltage (VH) thatis applied to the negative input terminal of the third comparator 306.During the seventh clock phase, the fourth predetermined voltage signal(VREF4) may be the predetermined low voltage (VL) that is applied to thenegative input terminal of the fourth comparator 308. Based upon theoutput signals from the comparators 306, 308 the sub-ADC 400 maygenerate new values for the high, low, and mid-switch control signals(h, l, and m). The sub-ADC 400 also generates two raw digital bits atthe end of the seventh clock phase, which are sent to the digitalsection 220 of FIG. 2.

FIG. 14 is a simplified circuit diagram illustrating the configurationof the sub-ADC 400 and the MDAC 500 during the eighth clock phase of theexample 10-bit A/D conversion process. During the eighth clock phase,the op-amp 555 generates a new residual voltage feedback signal VR₆ fromthe previous residual voltage feedback signal VR₅ based on the controlsignals (h, l, m) from the previous seventh clock phase. The residualvoltage feedback signal VR₆ is held and sampled on to the capacitors 521and 523 as well as sampled by the comparators 310, 312 of the sub-ADC400. During the eighth clock phase, the fifth predetermined voltagesignal (VREF5) may be the predetermined high voltage (VH) that isapplied to the negative input terminal of the fifth comparator 310.During the eighth clock phase, the sixth predetermined voltage signal(VREF6) may be the predetermined low voltage (VL) that is applied to thenegative input terminal of the sixth comparator 312. Based upon theoutput signals from the comparators 310, 312 the sub-ADC 400 maygenerate new values for the high, low, and mid-switch control signals(h, l, and m). The sub-ADC 400 also generates two raw digital bits atthe end of the eighth clock phase, which are sent to the digital section220 of FIG. 2.

FIG. 15 is a simplified circuit diagram illustrating the configurationof the sub-ADC 400 and the MDAC 500 during the ninth clock phase of theexample 10-bit A/D conversion process. During the ninth clock phase, theop-amp 555 generates a new residual voltage feedback signal VR₇ from theprevious residual voltage feedback signal VR₆ based on the controlsignals (h, l, m) from the previous eighth clock phase. The residualvoltage feedback signal VR₇ is held and sampled on to the capacitors 527and 525 as well as sampled by the comparators 302, 304, 306, 308 of thesub-ADC 400. During the ninth clock phase, the first and thirdpredetermined voltage signals (VREF1, VREF3) may be the predeterminedhigh voltage (VH) that is applied to the negative input terminals of thefirst and third comparators 302, 306. During the ninth clock phase, thesecond and fourth predetermined voltage signals (VREF2, VREF4) may bethe predetermined low voltage (VL) that is applied to the negative inputterminals of the second and fourth comparators 304, 308. Based upon theoutput signals from the comparators 302, 304, 306, 308 the sub-ADC 400may generate new values for the high, low, and mid-switch controlsignals (h, l, and m). The sub-ADC 400 also generates two raw digitalbits at the end of the ninth clock phase, which are sent to the digitalsection 220 of FIG. 2.

As explained above, the raw digital bits obtained from the sub-ADC 400in the example 10-bit A/D conversion were sent to the digital section220 of FIG. 2 during the first clock phase and during the third throughninth clock phases. In particular, the raw digital bits are sent to thealignment and synchronization block 230 of FIG. 2, where they arealigned and synchronized. During the tenth clock phase of the example10-bit A/D conversion process, a digital correction is performed in thecorrection block 240 to produce a 10-bit binary word at the end of thetenth clock phase, completing the example 10-bit A/D conversion. Theprocess may then be repeated in the manner described above to produce asecond 10-bit binary word. An observant reader will recognize that thenumber of raw digital bits obtained from the sub-ADC 400 in the example10-bit A/D conversion described above was actually greater than 10 bits.This discrepancy is accounted for because in each one of the clockphases that results in raw digital bits being output from the sub-ADC400, one of the raw bits is redundant and is discarded during furtherprocessing in the digital section 220. Thus, three raw digital bits fromthe sub-ADC during one clock phase produced two bits for the 10-bit A/Dconversion, and two raw digital bits from the sub-ADC during one clockphase produced one bit of the 10-bit A/D conversion.

According to the example 10-bit A/D conversion described above, fiveclock cycles are needed to produce a 10-bit binary word. Thus,generalizing to any n-bit A/D conversion where n is even, an n-bitbinary word may be produced in n/2 clock cycles. In alternativeembodiments, the sub-ADC 400 and the MDAC 500 could be configured toproduce two raw digital bits during, for example, the ninth clock phasethat was described above as producing three raw digital bits for theexample 10-bit A/D conversion. Thus, generalizing to any n-bit A/Dconversion where n is odd, an n-bit binary word may be produced in(n+1)/2 clock cycles. The number of clock cycles used in the example10-bit conversion described above is not significantly different fromthe number of clock cycles required by the single multi-bit A/Dconverter described in U.S. Pat. No. 6,535,157, which may produce twodigital bits during one clock phase of every clock cycle. However, thoseskilled in the art will appreciate that because the sub-ADC 400 and theMDAC 500 of the example embodiment can be continually reconfigured toproduce two raw digital bits during every clock phase for a clock cycleafter an initial clock cycle, the example embodiment can achieve thesame performance with reduced thermal noise, area, and power.

For example, in the 10-bit conversion described above, the first clockphase of the first clock cycle and the ninth clock phase of the fifthclock cycle were used to produce three raw digital bits from the sub-ADC400. The sub-ADC 400 was not used during the second clock phase of thefirst clock cycle. In the second through fourth clock cycles, however,by efficiently reconfiguring circuitry in the MDAC 500 during each clockphase to perform a different function, the sub-ADC 400 was used duringeach clock phase to produce two raw digital bits in each clock phase.Thus, according to example embodiments a single RSD A/D conversion stagecan be initially configured to output at least three raw bits during aninitial conversion clock cycle, then be subsequently reconfigured tooutput two raw bits during every clock phase of subsequent conversionclock cycles in order to determine the remaining bits of the A/Dconversion with reduced capacitance, reduced area, and reduced powerrequirements.

Based on the above, it should be apparent that example embodimentsinclude a single RSD stage that can be selectively reconfigured to havedifferent bit resolutions during different clock phases or clock cyclesof an A/D conversion process. In the particular example described above,the initial resolution was 2.5 bits, and the subsequent resolution was1.5 bits.

The example embodiment described above can achieve the same sample rateand resolution as the architecture described in U.S. Pat. No. 6,535,157,but the reconfiguration from a 2.5 bit resolution stage in the firstconversion clock cycle to a 1.5 bit resolution stage in subsequent clockcycles as described above enables it to do so with approximately a 40%reduction in total capacitance due to reduced thermal noise andapproximately a 25% reduction in area and power.

FIG. 16 is a flowchart illustrating a few example processes included ina method according to an example embodiment. Referring to FIG. 16, amethod 1600 according to an example embodiment starts with process 1610.Process 1610 includes generating, with a single RSD stage, at leastthree bits of raw digital data during a first clock phase of an A/Dconversion of an analog signal. Process 1620 occurs after process 1610,and includes generating, with the same single RSD stage, two bits ofdigital data during a second clock phase of the A/D conversion.

While the order of processes 1610 and 1620 as illustrated in FIG. 16 ispreferred, alternative embodiments may instead reverse the order,placing process 1620 prior to process 1610. In alternative embodiments,there may also be at least one intervening clock phase between the firstclock phase and the second clock phase of the A/D conversion of theanalog signal. In other words, the second clock phase of process 1620does not necessarily occur immediately after the first clock phase ofprocess 1610. It should also be appreciated that the first clock phaseof process 1610 need not be the sequentially first clock phase in anyparticular A/D conversion process, although this is preferred.

According to an example embodiment, a converter adapted to convert ananalog input signal into a digital output signal includes an analoginput terminal for receiving the analog input signal, and a RedundantSigned Digit (RSD) stage coupled to the analog input terminal. Accordingto the example embodiment, the RSD is stage is configured to receive theanalog input signal at the analog input terminal, produce a first numberof bits at a digital output from the analog input signal during a firsthalf of a first clock cycle, provide a residual feedback signal of theanalog input signal at the analog input terminal during a second half ofthe first clock cycle, and produce a second number of bits at thedigital output from the residual feedback signal during a first half ofa second clock cycle, the second number of bits less than the firstnumber of bits. According to the example embodiment, the converterfurther includes a digital section coupled to the digital output, thedigital section configured to perform a digital alignment and correctionon the first number of bits and the second number of bits to generatethe digital output signal.

According to an example embodiment, the RSD stage includes a MultiplyingDigital to Analog Converter (MDAC) that is operable to produce theresidual feedback signal, and a subsidiary Analog to Digital Converter(sub-ADC) that is operable to produce the first number of bits basedupon the analog input signal and that is operable to produce the secondnumber of bits based upon the residual feedback signal. According to anexample embodiment, the sub-ADC includes a plurality of comparatorscoupled to the analog input terminal and configured to compare theresidual feedback signal to a plurality of predetermined voltages, and alogic circuit coupled to the plurality of comparators and configured togenerate the first number of bits based at least upon outputs from afirst set of the comparators, the logic circuit further configured togenerate the second number of bits based at least upon outputs from asecond set of comparators, the second set of comparators a subset of thefirst set of comparators.

According to an example embodiment, the MDAC includes an operationalamplifier operable to generate the residual feedback signal, a firstcapacitor that is coupled to a first node, a second capacitor that iscoupled to the first node, a third capacitor that is coupled to a secondnode, a fourth capacitor that is coupled to the second node, a firstswitch that is coupled between an input of the op-amp and the firstnode, and a second switch that is coupled between the input of theop-amp and the second node. According to an example embodiment, the MDACfurther includes a third switch coupled between the first node and thesecond node.

According to an example embodiment, the RSD stage is further configuredto produce a third number of bits during a second half of the secondclock cycle, the third number equal to the second number. According toan example embodiment, the first number is three and the second numberis two.

According to another example embodiment, a cyclic Redundant Signed Digit(RSD) Analog to Digital (A/D) converter includes an input terminal forreceiving an analog input signal, a first switch connected between theinput terminal and a first node, the first switch operable to apply theanalog input signal to the first node, a second switch connected betweenthe first node and a second node, the second switch operable to apply aresidual voltage feedback signal to the first node, the first switchoperable to be closed when the second switch is open, the second switchoperable to be closed when the first switch is open. According to theembodiment, the cyclic RSD A/D converter includes an operationalamplifier having an output terminal connected to the second node, theoperational amplifier operable to generate the residual voltage feedbacksignal and apply it to the second node, and comparators, each comparatorhaving a first input coupled to the first node and an output, each ofthe comparators operable to compare a selected one of the analog inputsignal and the residual voltage feedback signal to a predeterminedvoltage signal. The cyclic RSD A/D converter further includes a logiccircuit coupled to the outputs of the comparators, the logic circuitoperable to generate a first digital output signal during a first clockphase of an A/D conversion and operable to generate a second digitaloutput signal during a second clock phase of the A/D conversion, thefirst digital output signal based upon the outputs from a first set ofthe comparators, the second digital output signal based upon the outputsfrom a second set of the comparators.

According to an example embodiment, the first digital output signalincludes three digital bits, while the second digital output signalconsists of two digital bits. According to an example embodiment, thesecond clock phase is subsequent to the first clock phase, and there isat least one intervening clock phase between the first clock phase andthe second clock phase. According to an example embodiment, the secondset of comparators is a subset of the first set of comparators.According to an example embodiment, the first digital output signalcomprises three digital bits, and the second digital output signalconsists of two digital bits. According to an example embodiment, aclock cycle of the A/D conversion consists of the first clock phase andthe second clock phase.

According to another example embodiment, a method for converting ananalog input signal into a plurality of digital bits during a pluralityof clock cycles using a single Redundant Signed Digit (RSD) stage of anAnalog to Digital (A/D) converter includes the steps of receiving theanalog input signal, and producing a first number of the digital bits ata first resolution during one of the clock cycles and producing a secondnumber of the digital bits at a second resolution during another one ofthe clock cycles.

According to an example embodiment, producing the first number ofdigital bits and the second number of digital bits comprises the stepsof producing the first number of the digital bits from the analog inputsignal during a first half of a first clock cycle, producing a firstresidual voltage from the analog input signal during a second half ofthe first clock cycle, and producing the second number of the digitalbits from the first residual voltage during a first half of a secondclock cycle. According to an example embodiment, the first resolution isat least 2.5 bits, and the second resolution is less than the firstresolution. According to an example embodiment, the second resolution is1.5 bits. According to an example embodiment, the first half of thesecond clock cycle occurs following the second half of the first clockcycle.

According to an example embodiment, the method further includes the stepof producing a second residual voltage from the first residual voltageduring a second half of the second clock cycle. According to an exampleembodiment, the method further includes the step of producing a thirdnumber of digital bits at the second resolution from the second residualvoltage during the second half of the second clock cycle.

It shall be apparent to those of ordinary skill, based upon the limitednumber of example embodiments described above, that many otherembodiments that incorporate one or more of the inventive principlesthat were associated with the described example embodiments exist. Inthe following paragraphs, more descriptions of example, non-limitingembodiments are presented.

While at least one example embodiment has been presented in theforegoing detailed description, it should be appreciated that a vastnumber of variations exist, especially with respect to choices of devicetypes and materials and the sequence of processes. It should further beemphasized that the example embodiments described above are onlyexamples, and are not intended to limit the scope, applicability, orconfiguration in any way. Rather, the detailed description of theexample embodiments provides those skilled in the art with a convenientroad map for implementing the inventive principles contained in theexample embodiments. The inventors regard the subject matter to includeall combinations and subcombinations of the various elements, features,functions and/or properties disclosed herein. It should also beunderstood that various changes can be made in the function andarrangement of elements without departing from the scope as set forth inthe appended claims and the legal equivalents thereof.

1. A converter adapted to convert an analog input signal into a digitaloutput signal, comprising: an analog input terminal for receiving theanalog input signal; a Redundant Signed Digit (RSD) stage coupled to theanalog input terminal, said RSD stage configured to: receive the analoginput signal at the analog input terminal; produce a first number ofbits at a digital output from the analog input signal during a firsthalf of a first clock cycle; provide a residual feedback signal of theanalog input signal at the analog input terminal during a second half ofthe first clock cycle; and produce a second number of bits at thedigital output from the residual feedback signal during a first half ofa second clock cycle, the second number of bits less than the firstnumber of bits; and a digital section coupled to the digital output, thedigital section configured to perform a digital alignment and correctionon the first number of bits and the second number of bits to generatethe digital output signal.
 2. The converter of claim 1, the RSD stagecomprising: A Multiplying Digital to Analog Converter (MDAC) that isoperable to produce the residual feedback signal; and A subsidiaryAnalog to Digital Converter (sub-ADC) that is operable to produce thefirst number of bits based upon the analog input signal and that isoperable to produce the second number of bits based upon the residualfeedback signal.
 3. The converter of claim 3, the sub-ADC comprising: aplurality of comparators coupled to the analog input terminal andconfigured to compare the residual feedback signal to a plurality ofpredetermined voltages; and a logic circuit coupled to the plurality ofcomparators and configured to generate the first number of bits based atleast upon outputs from a first set of the comparators, the logiccircuit further configured to generate the second number of bits basedat least upon outputs from a second set of comparators, the second setof comparators a subset of the first set of comparators.
 4. Theconverter of claim 3, the MDAC comprising: an operational amplifieroperable to generate the residual feedback signal; a first capacitorthat is coupled to a first node; a second capacitor that is coupled tothe first node; a third capacitor that is coupled to a second node; afourth capacitor that is coupled to the second node; a first switch thatis coupled between an input of the op-amp and the first node; and asecond switch that is coupled between the input of the op-amp and thesecond node.
 5. The converter of claim 5, the MDAC further comprising athird switch coupled between the first node and the second node.
 6. Theconverter of claim 1, wherein the RSD stage is further configured toproduce a third number of bits during a second half of the second clockcycle, the third number equal to the second number.
 7. The converter ofclaim 6, wherein the first number is three and the second number is two.8. A cyclic Redundant Signed Digit (RSD) Analog to Digital (A/D)converter comprising: an input terminal for receiving an analog inputsignal; a first switch connected between the input terminal and a firstnode, the first switch operable to apply the analog input signal to thefirst node; a second switch connected between the first node and asecond node, the second switch operable to apply a residual voltagefeedback signal to the first node, the first switch operable to beclosed when the second switch is open, the second switch operable to beclosed when the first switch is open; an operational amplifier having anoutput terminal connected to the second node, the operational amplifieroperable to generate the residual voltage feedback signal and apply itto the second node; comparators, each comparator having a first inputcoupled to the first node and an output, each of the comparatorsoperable to compare a selected one of the analog input signal and theresidual voltage feedback signal to a predetermined voltage signal; anda logic circuit coupled to the outputs of the comparators, the logiccircuit operable to generate a first digital output signal during afirst clock phase of an A/D conversion and operable to generate a seconddigital output signal during a second clock phase of the A/D conversion,the first digital output signal based upon the outputs from a first setof the comparators, the second digital output signal based upon theoutputs from a second set of the comparators.
 9. The cyclic RSD A/Dconverter of claim 8, the first digital output signal comprising threedigital bits, the second digital output signal consisting of two digitalbits.
 10. The cyclic RSD A/D converter of claim 9, wherein the secondclock phase is subsequent to the first clock phase, and wherein there isat least one intervening clock phase between the first clock phase andthe second clock phase.
 11. The cyclic RSD A/D converter of claim 9,wherein the second set of comparators is a subset of the first set ofcomparators.
 12. The cyclic RSD A/D converter of claim 8, the firstdigital output signal comprising three digital bits, the second digitaloutput signal consisting of two digital bits.
 13. The cyclic RSD A/Dconverter of claim 12, wherein a clock cycle of the A/D conversionconsists of the first clock phase and the second clock phase.
 14. Amethod for converting an analog input signal into a plurality of digitalbits during a plurality of clock cycles using a single Redundant SignedDigit (RSD) stage of an Analog to Digital (A/D) converter, the methodcomprising the steps of: receiving the analog input signal; andproducing a first number of the digital bits at a first resolutionduring one of the clock cycles and producing a second number of thedigital bits at a second resolution during another one of the clockcycles.
 15. The method of claim 14, wherein producing the first numberof digital bits and the second number of digital bits comprises thesteps of: producing the first number of the digital bits from the analoginput signal during a first half of a first clock cycle; producing afirst residual voltage from the analog input signal during a second halfof the first clock cycle; and producing the second number of the digitalbits from the first residual voltage during a first half of a secondclock cycle.
 16. The method of claim 14, wherein the first resolution isat least 2.5 bits, and wherein the second resolution is less than thefirst resolution.
 17. The method of claim 16, wherein the secondresolution is 1.5 bits.
 18. The method of claim 15, wherein the firsthalf of the second clock cycle occurs following the second half of thefirst clock cycle.
 19. The method of claim 15, further comprising thestep of producing a second residual voltage from the first residualvoltage during a second half of the second clock cycle.
 20. The methodof claim 19, further comprising the step of producing a third number ofdigital bits at the second resolution from the second residual voltageduring the second half of the second clock cycle.